TPS61040中文资料

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

LOW POWER DC/DC BOOST CONVERTER IN SOT-23 PACKAGE

FEATURESDESCRIPTION

D

1.8 V to 6 V Input Voltage Range

D

Adjustable Output Voltage Range up to 28 V

D

400 mA (TPS61040) and 250 mA (TPS61041)

D

D

D

D

D

Internal Switch Current

Up to 1 MHz Switching Frequency

28 µA Typical No Load Quiescent Current

1 µA Typical Shutdown Current

Internal Softstart

Available in a Tiny 5-Pin SOT23 Package

The TPS61040/41 is a high-frequency boost converter

dedicated for small to medium LCD bias supply and

white LED backlight supplies. The device is ideal to

generate output voltages up to 28 V from a dual cell

NiMH/NiCd or a single cell Li-Ion battery. The part can

also be ud to generate standard 3.3 V/5 V to 12 V

power conversions.

The TPS61040/41 operates with a switching frequency

up to 1 MHz. This allows the u of small external

components using ceramic as well as tantalum output

capacitors. Together with the tiny SOT23 package, the

TPS61040/41 gives a very small overall solution size.

The TPS61040 has an internal 400 mA switch current

limit, while the TPS61041 has a 250 mA switch current

limit, offering lower output voltage ripple and allows the

u of a smaller form factor inductor for lower power

applications. The low quiescent current (typically

28 µA) together with an optimized control scheme,

allows device operation at very high efficiencies over

the entire load current range.

APPLICATIONS

D

LCD Bias Supply

D

White-LED Supply for LCD Backlights

D

Digital Still Camera

D

PDAs, Organizers and Handheld PCs

D

Cellular Phones

D

Internet Audio Player

D

Standard 3.3 V/5 V to 12 V Conversion

TYPICAL APPLICATION

L1

10 µH

90

EFFICIENCY

vs

OUTPUT CURRENT

D1

V

OUT

V to 28 V

IN

E

f

f

i

c

i

e

n

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

DBV PACKAGE

(TOP VIEW)

SW

GND

FB

1

2

3

5

V

IN

4

EN

Ordering Information

†

TSWITCH CURRENT LIMITSOT23 PACKAGE (DBV)PACKAGE MARKING

A

–4040 to 85to85°C

400 mATPS61040DBVPHOI

250 mATPS61041DBVPHPI

†

The DBV package is available in tape & reel. Add “R” suffix (DBVR) to order quantities of 3000 parts.

functional block diagram

SW

Under Voltage

Lockout

Bias Supply

VIN

400 ns Min

Off Time

Error Comparator

FB

–

+

V = 1.233 V

REF

S

RS Latch

Logic

R

Current Limit

Gate

Driver

Power MOSFET

N-Channel

EN

6 µs Max

On Time

Soft

Start

+

_

R

SENSE

GND

Terminal Functions

TERMINAL

NAMENO.

SW1IConnect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to the drain of the internal

GND2Ground

FB3IThis is the feedback pin of the device. Connect this pin to the external voltage divider to program the desired output

EN4IThis is the enable pin of the device. Pulling this pin to ground forces the device into shutdown mode reducing the supply

VIN5ISupply voltage pin

I/O

DESCRIPTION

power MOSFET.

voltage.

current to less than 1 µA. This pin should not be left floating and needs to be terminated.

2

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

detailed description

operation

The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up

to 28 V. The device operates in a pul frequency modulation (PFM) scheme with constant peak current control.

This control scheme maintains high efficiency over the entire load current range, and with a switching frequency

up to 1 MHz, the device enables the u of very small external components.

The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference

voltage of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon

as the inductor current reaches the internally t peak current of typically 400 mA (TPS61040) or 250 mA

(TPS61041). Refer to the ction peak current control for more information. The cond criteria that turns off

the switch is the maximum on-time of 6 µs (typical). This is just to limit the maximum on-time of the converter

to cover for extreme conditions. As the switch is turned off the external Schottky diode is forward biad

delivering the current to the output. The switch remains off for a minimum of 400 ns (typical), or until the feedback

voltage drops below the reference voltage again. Using this PFM peak current control scheme the converter

operates in discontinuous conduction mode (DCM) where the switching frequency depends on the output

current, which results in very high efficiency over the entire load current range. This regulation scheme is

inherently stable, allowing a wider lection range for the inductor and output capacitor.

peak current control

The internal switch turns on until the inductor current reaches the typical dc current limit (I

LIM

) of 400 mA

(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, the actual current

exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:

) 100ns+I

Vin

peak(typ)LIM

L

I

+400mA)

Vin

100nsfortheTPS61040

peak(typ)

L

I

+250mA)

Vin

100nsfortheTPS61041

peak(typ)

L

I

The higher the input voltage and the lower the inductor value, the greater the peak.

By lecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current

limit requirements. A lower current limit supports applications requiring lower output power and allows the u

of an inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output

voltage ripple as well.

softstart

All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This

can cau voltage drops at the input rail during start up and may result in an unwanted or early system shut

down.

The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from

256 cycles to

I

LIM

4

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

detailed description (continued)

enable

Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 µA (typical). Since

there is a conductive path from the input to the output through the inductor and Schottky diode, the output

voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should not

be left floating. Using a small external transistor disconnects the input from the output during shutdown as shown

in Figure 18.

undervoltage lockout

An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the

input voltage is below the undervoltage threshold the main switch is turned off.

absolute maximum ratings over operating free-air temperature (unless otherwi noted)

†

Supply voltages on pin VIN (e Note 1). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 7 V

Voltages on pins EN, FB (e Note 1). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to V + 0.3 V

IN

Switch voltage on pin SW (e Note 1). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 V

Continuous power dissipation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table

Operating junction temperature, T

J

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 150°C

Storage temperature, T

Stg

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C

Lead temperature (soldering 10 conds). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C

†

Stress beyond tho listed under “absolute maximum ratings” may cau permanent damage to the device. The are stress ratings only, and

functional operation of the device at the or any other conditions beyond tho indicated under “recommended operating conditions” is not

implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

NOTE 1:All voltage values are with respect to network ground terminal.

DISSIPATION RATING TABLE

PACKAGE

DBV357 mW3.5 mW/°C192 mW140 mW

T ≤ 25°CDERATING FACTORT = 70°CT = 85°C

AAA

POWER RATINGABOVE T = 25°CPOWER RATINGPOWER RATING

A

NOTE:The thermal resistance junction to ambient of the 5-pin SOT23 is 250°C/W.

recommended operating conditions

MINTYPMAXUNIT

Input voltage range, Vin1.86V

Output voltage range, V28V

OUT

Inductor (e Note 2), L2.210µH

Switching frequency (e Note 2), f1MHz

Input capacitor (e Note 2), C4.7µF

in

Output capacitor (e Note 2), C1µF

OUT

Operating ambient temperature, T–4085°C

A

Operating junction temperature, T–40125°C

J

NOTE 2:Refer to application ction for further information

4

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

electrical characteristics, Vin = 2.4 V, EN = VIN, T= –40°C to 85°C typical values are at T= 25°C

A,A

(unless otherwi noted)

supply current

PARAMETERTEST CONDITIONSMINTYPMAXUNIT

VinInput voltage range1.86V

IOperating quiescent currentI = 0 mA, not switching, V = 1.3 V2850µA

QOUTFB

IShutdown currentEN=GND0.11µA

SD

VUnder-voltage lockout threshold1.51.7V

UVLO

enable

PARAMETERTEST CONDITIONSMINTYPMAXUNIT

VEN high level input voltage1.3V

IH

VEN low level input voltage0.4V

IL

IEN input leakage currentEN = GND or VIN0.11µA

I

power switch and current limit

PARAMETERTEST CONDITIONSMINTYPMAXUNIT

VswMaximum switch voltage30V

tMinimum off time250400550ns

off

tMaximum on time467.5µs

on

RMOSFET on-resistanceVin = 2.4 V; Isw = 200 mA; TPS610406001000mΩ

DS(ON)

RMOSFET on-resistanceVin = 2.4 V; Isw = 200 mA; TPS610417501250mΩ

DS(ON)

IMOSFET current limitTPS61040350400450mA

LIM

IMOSFET current limitTPS61041215250285mA

LIM

MOSFET leakage currentV = 28 V110µA

sw

output

PARAMETERTEST CONDITIONSMINTYPMAXUNIT

VAdjustable output voltage rangeVin28V

OUT

VInternal voltage reference1.233V

ref

IFeedback input bias currentV = 1.3 V1µA

FBFB

VFeedback trip point voltage1.8 V ≤ Vin ≤ 6.0 V1.2081.2331.258V

FB

Line regulation (e Note 3)0.05%/V

Load regulation (e Note 3)Vin = 2.4 V; Vout = 18 V; 0 mA ≤ Iout ≤ 30 mA0.15%/mA

1.8 V ≤ Vin ≤ 6.0 V; Vout = 18 V; Iload = 10 mA

Cff = not connected

NOTE 3:The line and load regulation depend on the external component lection. Refer to the application ction for further information.

5

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

TYPICAL CHARACTERISTICS

Table of Graphs

FIGURE

ηEfficiency

IQuiescent currentvs Input voltage and temperature5

Q

VFeedback voltagevs Temperature6

FB

ISwitch current limitvs Temperature7

SW

ISwitchcurrentlimitSwitch current limit

CL

RDSRDSon

on

Line transient respon12

Load transient respon13

Start-up behavior14

vs Load current1, 2, 3

vs Input voltage4

vs Supply voltage, TPS610418

vs Supply voltage, TPS610409

vs Temperature10

vs Supply voltage

11

EFFICIENCY

vs

OUTPUT CURRENT

90

88

86

84

E

f

f

i

c

i

e

n

c

y

–

%

82

80

78

76

74

72

70

0.10110100

I – Output Current – mA

O

V = 2.4 V

I

V = 3.6 V

I

E

f

f

i

c

i

e

n

c

y

–

%

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

TYPICAL CHARACTERISTICS

EFFICIENCY

vs

LOAD CURRENT

90

88

86

84

E

f

f

i

c

i

e

n

c

y

–

%

82

80

78

76

74

72

70

0.101

10

I – Load Current – mA

L

100

E

f

f

i

c

i

e

n

c

y

–

%

L = 3.3 µH

V = 18 V

O

L = 10 µH

90

88

86

84

82

80

78

76

74

72

70

123456

V – Input Voltage – V

I

L = 10 µH

V = 18 V

O

EFFICIENCY

vs

INPUT VOLTAGE

I = 10 mA

O

I = 5 mA

O

Figure 3Figure 4

TPS61040

QUIESCENT CURRENT

vs

INPUT VOLTAGE

40

35

Q

u

i

e

s

c

e

n

t

C

u

r

r

e

n

t

–

µ

A

30

25

T = –40°C

A

20

15

10

5

0

1.82.433.64.24.85.46

1.23

–40–20020406080100120

T = 85°C

A

1.238

T = 27°C

A

V

F

B

–

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

TYPICAL CHARACTERISTICS

TPS61040/41

TPS61041

SWITCH CURRENT LIMIT

vs

FREE-AIR TEMPERATURE

430

410

I

(

S

W

)

–

S

w

i

t

c

h

C

u

r

r

e

n

t

L

i

m

i

t

–

m

A

390

I

(

C

L

)

–

C

u

r

r

e

n

t

L

i

m

i

t

–

m

A

370

350

330

310

290

270

250

230

–40–30–20–100102030405060708090

T – Temperature – °C

A

TPS61041

242

240

1.82.433.64.24.85.46

TPS61040

260

258

256

254

252

250

248

246

244

CURRENT LIMIT

vs

SUPPLY VOLTAGE

T = 27°C

A

V – Supply Voltage – V

CC

Figure 7Figure 8

TPS61040

TPS61040/41

STATIC DRAIN-SOURCE ON-STATE RESISTANCE

vs

FREE-AIR TEMPERATURE

1200

420

415

I

(

C

L

)

–

C

u

r

r

e

n

t

L

i

m

i

t

–

m

A

410

405

400

395

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

TYPICAL CHARACTERISTICS

TPS61040/41

STATIC DRAIN-SOURCE ON-STATE RESISTANCE

vs

SUPPLY VOLTAGE

1000

900

800

TPS61041

700

600

500

400

300

200

100

0

1.82.433.64.24.85.46

TPS61040

r

D

S

(

o

n

)

–

S

t

a

t

i

c

D

r

a

i

n

-

S

o

u

r

c

e

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

TYPICAL CHARACTERISTICS

V = 18 V

O

V

O

100 mA/div

V

O

1 mA to 10 mA

200 µS/div

Figure 13. Load Transient Respon

V = 18 V

O

V

O

5 V/div

EN

1 V/div

I

I

50 mA/div

Figure 14. Start-Up Behavior

10

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

inductor lection, maximum load current

Since the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability

of the regulator. The lection of the inductor together with the nominal load current, input and output voltage

of the application determines the switching frequency of the converter. Depending on the application, inductor

values between 2.2µH up to 47 µH are recommended. The maximum inductor value is determined by the

maximum on time of the switch, typically 6 µs. The peak current limit of 400 mA/250mA (typically) should be

reached within this 6-µs period for proper operation.

The inductor value determines the maximum switching frequency of the converter. Therefore, lect the

inductor value that ensures the maximum switching frequency at the converter maximum load current is not

exceeded. The maximum switching frequency is calculated by the following formula:

fS

max

+

Where:

I = Peak current as described in the previous peak current control ction

P

L = Selected inductor value

Vin

min

= The highest switching frequency occurs at the minimum input voltage

If the lected inductor value does not exceed the maximum switching frequency of the converter, the next step

is to calculate the switching frequency at the nominal load current using the following formula:

fS

I

Where:

I = Peak current as described in the previous peak current control ction

P

L = Selected inductor value

I

load

= Nominal load current

Vd = Rectifier diode forward voltage (typically 0.3V)

A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.

The inductor value has less effect on the maximum available load current and is only of condary order. The

best way to calculate the maximum available load current under certain operating conditions is to estimate the

expected converter efficiency at the maximum load current. This number can be taken out of the efficiency

graphs shown in Figures 1, 2, 3, and 4. The maximum load current can then be estimated as follows:

2

L fS

max

P

I

+h

loadmax

2 (Vout*Vin)

Vin

(Vout–Vin)

min

I

L Vout

P

ǒ

load

Ǔ

+

2 I

load

(Vout–Vin)Vd)

I

P

2

L

I

Where:

I = Peak current as described in the previous peak current control ction

P

L = Selected inductor value

fS

max

= Maximum switching frequency as calculated previously

η = Expected converter efficiency. Typically 70% to 85%

11

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

The maximum load current of the conveter is the current at the operation point where the coverter starts to enter

the continuous conduction mode. Usually the converter should always operate in discontinuous conduction

mode.

Last, the lected inductor should have a saturation current that meets the maximum peak current of the

converter (as calculated in the peak current control ction). U the maximum value for I

LIM

for this calculation.

Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency

of the converter. Refer to the Table 1 and the typical applications for the inductor lection.

Table 1. Recommended Inductor for Typical LCD Bias Supply (e Figure 15)

DEVICEINDUCTOR VALUECOMPONENT SUPPLIERCOMMENTS

10 µHSumida CR32-100High efficiency

10 µHSumida CDRH3D16-100High efficiency

TPS61040

10 µHMurata LQH4C100K04High efficiency

4.7 µHSumida CDRH3D16-4R7Small solution size

4.7 µHMurata LQH3C4R7M24Small solution size

TPS6104110 µHMurata LQH3C100K24

High efficiency

Small solution size

tting the output voltage

The output voltage is calculated as:

V1)

out

+1.233V

R1

R2

ǒǓ

For battery powered applications a high impedance voltage divider should be ud with a typical value for R2

of ≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be ud to reduce the noi nsitivity

of the feedback pin.

A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the

error comparator. Without a feedforward capacitor, or one who value is too small, the TPS61040/41 shows

double puls or a pul burst instead of single puls at the switch node (SW), causing higher output voltage

ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.

The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good

starting point is to u a 10 pF feedforward capacitor. As a first estimation, the required value for the feedforward

capacitor at the operation point can also be calculated using the following formula:

C

FF

+

1

2 p

fS

R1

20

Where:

R1 = Upper resistor of voltage divider

fS = Switching frequency of the converter at the nominal load current (See previous ction for calculating the

switching frequency)

C

FF

= Choo a value that comes clost to the result of the calculation

12

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

The larger the feedforward capacitor the wor the line regulation of the device. Therefore, when concern for

line regulation is paramount, the lected feedforward capacitor should be as small as possible. See the next

ction for more information about line and load regulation.

line and load regulation

The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV

peak-to-peak voltage ripple on the feedback pin FB gives good results.

Some applications require a very tight line regulation and can only allow a small change in output voltage over

a certain input voltage range. If no feedforward capacitor C is ud across the upper resistor of the voltage

FF

feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage

ripple is higher becau the TPS61040/41 shows output voltage bursts instead of single puls on the switch

pin (SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage

ripple.

If a larger output capacitor value is not an option, a feedforward capacitor C

FF

can be ud as described in the

previous ction. The u of a feedforward capacitor increas the amount of voltage ripple prent on the

feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the wor the line regulation.

There are two ways to improve the line regulation further:

1.U a smaller inductor value to increa the switching frequency which will lower the output voltage ripple,

as well as the voltage ripple on the feedback pin.

2.Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback

pin down to 50 mV again. As a starting point, the same capacitor value as lected for the feedforward

capacitor C

FF

can be ud.

output capacitor lection

For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low

ESR value but tantalum capacitors can be ud as well, depending on the application.

Assuming the converter does not show double puls or pul bursts on the switch node (SW), the output

voltage ripple can be calculated as:

I

L

I

1

P

DV+ )I ESR

out

out

–

P

C

out

fS(Iout)

Vout)Vd–Vin

Where:

I = Peak current as described in the previous peak current control ction

P

L = Selected inductor value

I

out

= Nominal load current

fS (I

out

) = Switching frequency at the nominal load current as calculated previously

Vd = Rectifier diode forward voltage (typically 0.3V)

C

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

Table 2. Recommended Input and Output Capacitors

DEVICECAPACITORVOLTAGE RATINGCOMPONENT SUPPLIERCOMMENTS

4.7 µF/X5R/08056.3 VTayo Yuden JMK212BY475MGC/C

10 µF/X5R/08056.3 VTayo Yuden JMK212BJ106MGC/C

TPS61040/41

1.0 µF/X7R/120625 VTayo Yuden TMK316BJ105KLC

1.0 µF/X5R/120635 VTayo Yuden GMK316BJ105KLC

4.7 µF/X5R/121025 VTayo Yuden TMK325BJ475MGC

INOUT

INOUT

OUT

OUT

OUT

input capacitor lection

For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 µF ceramic input

capacitor is sufficient for most of the applications. For better input voltage filtering this value can be incread.

Refer to Table 2 and typical applications for input capacitor recommendations.

diode lection

To achieve high efficiency a Schottky diode should be ud. The current rating of the diode should meet the peak

current rating of the converter as it is calculated in the ction peak current control. U the maximum value

for I

LIM

for this calculation. Refer to Table 3 and the typical applications for the lection of the Schottky diode.

Table 3. Recommended Schottky Diode for Typical LCD Bias Supply (e Figure 15)

DEVICEREVERSE VOLTAGECOMPONENT SUPPLIERCOMMENTS

30 VON Semiconductor MBR0530

TPS61040/41

20 VON Semiconductor MBR0520

20 VON Semiconductor MBRM120LHigh efficiency

30 VToshiba CRS02

layout considerations

Typical for all switching power supplies, the layout is an important step in the design; especially at high peak

currents and switching frequencies. If the layout is not carefully done, the regulator might show noi problems

and duty cycle jitter.

The input capacitor should be placed as clo as possible to the input pin for good input voltage viltering. The

inductor and diode should be placed as clo as possible to the switch pin to minimize the noi coupling into

other circuits. Since the feedback pin and network is a high impedance circit the feedback network should be

routed away from the inductor. The feedback pin and feedback network should be shielded with a ground plane

or trace to minimize noi coupling into this circuit.

Wide traces should be ud for connections in bold as shown in Figure 15. A star ground connection or ground

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

L1

D1

V

O

C

FF

C

O

V

IN

C

IN

V

IN

SW

FB

R1

EN

GND

R2

Figure 15. Layout Diagram

L1

10 µH

TPS61040

V

IN

C1

4.7 µF

SW

FB

EN

GND

R2

160 kΩ

R1

2.2 MΩ

D1

V

IN

1.8 V to 6 V

V

OUT

18 V

C

FF

22 pF

C2

1 µF

L1:Sumida CR32–100

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2:Tayo Yuden TMK316BJ105KL

Figure 16. LCD Bias Supply

L1

10 µH

TPS61040

V

IN

1.8 V to 6 V

C1

4.7 µF

V

IN

SW

FB

EN

GND

R2

160 kΩ

R1

2.2 MΩ

D1

V

O

18 V

C

FF

22 pF

C2

1 µF

DAC or Analog Voltage

0 V = 25 V

1.233 V = 18 V

L1:Sumida CR32-100

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2:Tayo Yuden GMK316BJ105KL

Figure 17. LCD Bias Supply With Adjustable Output Voltage

15

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

R3

200 kΩ

L1

10 µH

TPS61040

V

IN

C1

4.7 µF

SW

FB

EN

GND

BC857C

D1

V

OUT

18 V / 10 mA

R1

2.2 MΩ

C2C3

1 µF0.1 µF

R2

160 kΩ

C

FF

22 pF

(Optional)

L1:Sumida CR32-100

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2:Tayo Yuden TMK316BJ105KL

V

IN

1.8 V to 6 V

Figure 18. LCD Bias Supply With Load Disconnect

D3

V2 = –10 V/15 mA

D2

L1

6.8 µH

TPS61040

V = 2.7 V to 5 V

IN

C1

4.7 µF

V

IN

SW

FB

EN

GND

R2

210 kΩ

R1

1.5 MΩ

C3

1 µF

D1

V1 = 10 V/15 mA

C

FF

22 pF

C2

1 µF

L1:Murata LQH4C6R8M04

D1, D2, D3:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2, C3, C4:Tayo Yuden EMK316BJ105KF

C4

4.7 µF

Figure 19. Positive and Negative Output LCD Bias Supply

16

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

APPLICATION INFORMATION

L1

6.8 µH

TPS61040

V 3.3 V

IN

C1

10 µF

V

IN

SW

FB

EN

GND

R2

205 kΩ

R1

1.8 MΩ

D1

V 12 V/35 mA

O=

C

FF

4.7 pF

C2

4.7 µF

L1:Murata LQH4C6R8M04

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BJ106MG

C2:Tayo Yuden EMK316BJ475ML

Figure 20. Standard 3.3-V to 12-V Supply

3.3 µH

TPS61040

1.8 V to 4 VSW

V

IN

FB

EN

GND

R2

200 kΩ

R1

620 kΩ

D1

5 V/45 mA

C

FF

3.3 pF

C2

4.7 µF

L1:Murata LQH4C3R3M04

D1:Motorola MBR0530

C1, C2:Tayo Yuden JMK212BY475MG

C1

4.7 µF

Figure 21. Dual Battery Cell to 5V/50mA Conversion Efficiency≈ 84% at Vin = 2.4 V to Vo = 5 V/45 mA

L1

10 µH

D1

V = 2.7 V to 6 V

CC

C1

4.7 µF

PWM

100 Hz to 500 Hz

V

IN

SW

FB

D2

24 V

(Optional)

C2

1 µF

L1:Murata LQH4C100K04

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2:Tayo Yuden TMK316BJ105KL

EN

GND

R

S

82 Ω

Figure 22. White LED Supply With Adjustable Brightness Control Using a PWM Signal on the Enable Pin

Efficiency≈ 86% at Vin = 3 V, I = 15 mA

LED

17

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

L1

10 µH

D1

MBRM120L

V = 2.7 V to 6 V

CC

V

IN

SW

FB

D2

24 V

(Optional)

C2

†

100 nF

C1

4.7 µF

EN

GND

R1

120 kΩ

R

S

110 Ω

Analog Brightness Control

3.3 V≅ Led Off

0 V≅ Iled = 20 mA

R2 160 kΩ

L1:Murata LQH4C3R3M04

D1:Motorola MBR0530

C1:Tayo Yuden JMK212BY475MG

C2:Standard Ceramic Capacitor

†

A smaller output capacitor value for C2 will cau a larger LED ripple

Figure 23. White LED Supply With Adjustable Brightness Control

Using an Analog Signal on the Feedback Pin

18

TPS61040

TPS61041

SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002

MECHANICAL DATA

DBV (R-PDSO-G5)PLASTIC SMALL-OUTLINE

0,95

54

0,50

0,30

0,20

M

1,70

1,50

3,00

2,60

0,15 NOM

13

3,00

2,80

Gage Plane

0,25

0°–8°

0,55

0,35

Seating Plane

1,45

0,95

0,05 MIN

0,10

4073253-4/F 10/00

NOTES:A.All linear dimensions are in millimeters.

B.This drawing is subject to change without notice.

C.Body dimensions do not include mold flash or protrusion.

D.Falls within JEDEC MO-178.

19

IMPORTANT NOTICE

Texas Instruments Incorporated and its subsidiaries (TI) rerve the right to make corrections, modifications,

enhancements, improvements, and other changes to its products and rvices at any time and to discontinue

any product or rvice without notice. Customers should obtain the latest relevant information before placing

orders and should verify that such information is current and complete. All products are sold subject to TI’s terms

and conditions of sale supplied at the time of order acknowledgment.